Dc current cancellation scheme for an optical receiver

ABSTRACT

In high data rate receivers, comprising a photodetector (PD) and a transimpedance amplifier (TIA), a transmitted optical signal typically has poor extinction ratio, which translates into a small modulated current with a large DC current at the output of the PD. The large DC current saturates the TIA, which significantly degrades the gain and bandwidth performance. Accordingly, cancelling photo diode DC current in high data rate receivers is important for proper receiver operation. A DC current cancellation loop, comprising a low pass filter section and a trans-conductance cell (GM) are connected to the input of the TIA. PD DC current I DC  is drawn from the input node of the TIA in the GM cell, such that the cancellation loop maintains the DC voltage value of the TIA input node to be the same as a reference voltage (V REF ).

TECHNICAL FIELD

The present invention relates to an optical receiver, and in particularto an optical receiver with DC current cancellation.

BACKGROUND

A typical optical receiver front-end is composed of a photo diode (PD) 1followed by a trans-impedance amplifier (TIA) 2 and main amplifiers(MAs) 3 as shown in FIG. 1. The PD 1 receives a transmitted opticalsignal 4 and generates a current 6 proportional to the received opticalpower of the received optical signal 4. The ratio between PD outputcurrent 6 to the input optical power of the optical signal 4 is thephoto diode responsivity (R). The TIA 2 converts the PD current 6 tovoltage, which is then amplified by MAs 3 to the desired signal levelfor the decision circuitry 7. For NRZ modulation, the modulation depthof the optical signal 4 is defined by its extinction ratio, which is theratio between optical power for symbol one (P₁) and optical power forsymbol zero (P₀). In high data rate receivers, the transmitted opticalsignal 4 has poor extinction ratio and translates into a small modulatedcurrent with a large DC current at the output of the photo diode 1. Thelarge DC current saturates the receiver front-end (TIA 2 and MAs 3) andsignificantly degrades the gain and bandwidth performance. Consequently,cancelling photo diode DC current in high data rate receivers is desiredfor proper receiver operation, i.e. to have zero average modulated PDcurrent 6.

Moreover, for coherent optical communication links, mixing local laserpower and the received modulated optical signal 4 using the photo diode1 results in very large DC current. PD DC current is expressed as,

I _(DC) =R×(P _(LO) +P _(S)),   (1)

where P_(LO) is the local optical laser power and P_(S) is the receivedoptical signal power. Equation (1) shows that the photo diode output DCcurrent of the PD 1 in coherent optical communication links depends onthe local laser power and the optical received power. For example, aphoto diode 1 with responsivity (R) of 1 A/W results in 4 mA DC currentat 6 dBm local laser power input. Such a large DC current is more thanenough to saturate the receiver front-end and severely degrades itsperformance. Thus, it is very important to have DC current cancellationcircuitry in front of the TIA 2 of coherent optical communication links.

FIG. 2 shows a conventional way to AC couple receiver photo diodes 11 tothe a front-end TIA 12 using passive AC coupling circuitry. An ACcoupling capacitor (C_(C)) is inserted between the photo diode 11 andthe front-end TIA 12 to block the DC current; however, it bypasses themodulated AC current to the TIA 12. A biasing resistor (R_(C)) is usedto bias the photo diode anode voltage to be reverse biased, and providesan alternative path for the photo diode DC current I_(DC). The biasingresistor R_(C) with the AC coupling capacitor C_(C) forms a high passfilter section in the RF signal path and its cutoff frequency (FC) iscalculated as,

$\begin{matrix}{F_{C} = \frac{1}{2\pi \; R_{C}C_{C}}} & (2)\end{matrix}$

Typically, the required TIA low cutoff frequency (FC) is around 100 kHzwhich requires either large AC coupling capacitor C_(C) or huge biasingresistor R_(C). As an example, a capacitor C_(C) with a capacitance ofat least 1.6 pF with a resistor R_(C) with a resistance of at least 1 MΩare required to achieve cutoff frequency of 100 kHz. Yet, this techniquesuffers from two main drawbacks: 1) C_(C) parasitic capacitance, and 2)photo diode biasing. For bulk silicon technologies, the bottom plateground parasitic capacitance of the coupling capacitor C_(C) is around10% of its value and degrades the front-end TIA bandwidth, which isdefined by its input node capacitance. Thus, there is a maximum couplingcapacitor (C_(C)) that can be used without degrading the TIA bandwidth.On the other hand, the biasing voltage across the photo diode 11 isdefined by the following equation:

V _(BIAS) =V _(PD) −V _(B) −I _(DC) ×R _(C),   (3)

where V_(BIAS) is the reverse biasing voltage across the photo diode PNjunction. High photo diode reverse biasing voltage is required to obtaingood photo diode responsivity and low PN junction capacitance. However,equation (3) shows that V_(BIAS) depends on PD average current and leadsto different PD biasing for different received optical power.Furthermore, a large R_(C) value impedes receiving high optical powerlevels as the DC current will be large and the voltage drop across R_(C)will be huge. As a numerical example, an I _(DC) of 10 μA leads to a 10Vdrop on a 1 MΩ resistor R_(C), which is not practical. Moreover, thesituation in coherent optical receivers is much worse as the photo diodeDC current is around 1 mA and requires an R_(C) of less than 1 kΩ forless than 1 V drop across the biasing resistor R_(C).

An object of the present invention is to overcome the shortcomings ofthe prior art by providing a DC current cancellation loop for use with afully differential front-end TIA structure.

SUMMARY OF THE INVENTION

Accordingly, the present invention relates to an optical receivercomprising:

a first photodetector (PD) for converting a first input optical signalinto a first PD current comprising a first AC component and a first DCcomponent;

a transimpendance amplifier (TIA) for converting the first AC componentinto a first voltage signal; and

a first DC cancellation loop including an input and an output betweenthe first PD and the TIA for cancelling the first DC component, thefirst DC cancellation loop comprising:

a first input and a first output connected to an input of the TIA;

a first trans-conductance cell (G_(M)), capable of drawing in the firstDC component, such that the first DC cancellation loop maintains a firstDC voltage value of the first input of the TIA the same as a firstreference voltage (V_(REF)1), which represents an actual TIA inputvoltage for a zero DC current condition; and

a first low pass filter.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described in greater detail with reference to theaccompanying drawings which represent preferred embodiments thereof,wherein:

FIG. 1 illustrates a schematic diagram of a conventional NRZ opticalreceiver;

FIG. 2 illustrates a schematic diagram of a conventional differentialoptical receiver;

FIG. 3 illustrates a schematic diagram of a fully differential opticalreceiver in accordance with an embodiment of the present invention;

FIG. 4 illustrates a photodetector and differential TIA structure of theoptical receiver of FIG. 3;

FIG. 5 illustrates an embodiment of a transconductance cell inaccordance with the present invention; and

FIG. 6 illustrates another embodiment of a photodetector and TIAstructure of an optical receiver of the present invention.

DETAILED DESCRIPTION

While the present teachings are described in conjunction with variousembodiments and examples, it is not intended that the present teachingsbe limited to such embodiments. On the contrary, the present teachingsencompass various alternatives and equivalents, as will be appreciatedby those of skill in the art.

FIG. 3 illustrates an embodiment of an electronic component in apackaged optical receiver 30 in accordance with an embodiment of thepresent invention. An input signal enters at a port 31. A small portionof the input optical signal, e.g. less than 5%, may be split off andsent to a monitor photodiode 32, which generates an electrical signalthat may be used to monitor properties of the input optical signal, suchas its power content. In other embodiments, the power of the inputoptical signal can be monitored using different hardware. The remainderof the input optical signal may be sent through a variable opticalattenuator 33, which can adjust the signal intensity. A polarizationbeam splitter (PBS) 34 splits the remainder of the input optical signalinto x-polarized (X-Pol) and y-polarized (Y-Pol) components. The X-Polcomponent is sent to a 90° hybrid mixer 36, and the Y-Pol component issent to a 90° hybrid mixer 37. Simultaneously, a local oscillator 38provides a signal that is split by a beam splitter 39, and components ofwhich are sent to each of 90° hybrid mixers 36 and 37. The 90° hybridmixers 36 and 37 are optical components that each generate two phasedifferentiated optical signals, the XI and XQ signals and the YI and YQsignals, respectively. Finally, each of the four phase differentiatedsignals are converted to electrical signals by respective opticalreceivers 40, including photodetectors 41 and transimpedance amplifiers(TIA) 42. The electrical signals are then provided at four respectiveoutput terminals, which may be single-sided signals referenced to acommon ground or may be differential signals.

FIG. 4 illustrates an embodiment of the optical receiver 40 including anactive AC coupling circuitry disposed between the photo diode (PD) 41and the front-end differential TIA 42. An analog DC cancellation loop 45a and 45 b is located prior to each input of the TIA 42, and draws thephoto diode DC current I_(DC), whereby only the AC signal I_(AC) iscoupled to the TIA 42. Two different analog cancellation loops 45 a and45 b are used at both TIA inputs. Each cancellation loop 45 a and 45 bcomprises a low pass filter section 46 a and 46 b, e.g. comprised of acancellation resistor R_(C) and a cancellation capacitor C_(C), and atrans-conductance cell (GM), such that the input and output of eachcancellation loop 45 a and 45 b are connected to the input of the TIAinput 42, i.e. both the input and output of the loops 45 a and 45 b arebetween the PD 41 and the TIA 42, as shown in FIG. 4. Photo diode DCcurrent I_(DC) is drawn from the input node of the TIA 42 in the G_(M)cell, such that the analog loops 45 a and 45 b maintain the DC voltagevalue of the TIA input node to be the same as a reference voltage(V_(REF)). This reference voltage V_(REF) represent the actual TIA inputnode voltage for a zero DC current condition, which means that onaverage there is no current flowing into each of the differential TIAinputs. The term “zero” DC current means very little current, e.g. lessthan 100 uA or, ideally, less than a value that would significantlyimpact other TIA performance characteristics, such as linearity andbandwidth. Accordingly, the DC cancellation loops 45 a and 45 b greatlypreserve the linearity of the TIA 42 and the optical receiver 40,because if the DC current is not canceled, it would flow into the TIA 42and change the bias point, output signal common mode, and may evencompletely saturate the TIA 42, rendering it useless.

A method of generating V_(REF) includes using a replica TIA 47, alsoshown in FIG. 4. The replica TIA 47 includes floating inputs and outputsand substantially the same structure as the main TIA 42, in order togenerate a correct V_(REF) voltage, corresponding to a zero DC inputcurrent condition. Using a replica TIA 47 provides a simple way to trackthe main TIA 42 across process, voltage and temperature (PVT)variations, and to automatically generate a correct V_(REF) in allcases. The trade-off of using the replica TIA 47 is additional powerdissipation. In order to minimize the additional power dissipation fromthe replica TIA 47, it may be scaled down. For example, the replica TIA47 may use a fraction, e.g. ¼, of the bias current of the main TIA 42,flowing through resistors RF that are 4× larger than the resistors inthe main TIA 42. Accordingly, the same V_(REF) voltage is generated,while burning only ¼ of the power of the main TIA 42. Scaling willreduce the quality of the PVT tracking of the replica TIA 47, so thescaling factor has to be selected carefully, to achieve the perfectbalance between PVT tracking and power dissipation. It should be notedthat V_(REF) may be generated in a variety of other ways, without usinga replica TIA. For example, a custom analog circuit may be constructedto simultaneously achieve the goals of correct V_(REF) generation, withPVT tracking, and minimum power dissipation.

There are two important specifications that are advantageous from the DCcancellation loops 45 a and 45 b that define their loop gain-bandwidthproduct (GBW). First, the cancellation loops 45 a or 45 b should notaffect or attenuate the received high speed signal. Second, the loops 45a or 45b should track any variation in the photo diode DC current I_(DC)and completely cancel it. This implies that the speed of the analogcancellation loops 45 a and 45 b are bonded by two main upper and lowerlimits, which are the lowest frequency component of the received data(upper limit) and the fastest variation of the photo diode DC currentI_(DC) (lower limit). The gain-bandwidth product of this analog loop 45a or 45 b is calculated as,

$\begin{matrix}{{{GBW} = {\frac{G_{M} \times R_{F}}{A_{O}} \cdot \frac{1}{2\pi \; R_{C}C_{C}}}};} & (5)\end{matrix}$

where R_(F) is the TIA feedback resistor and A_(O) is the TIAfeed-forward amplifier DC gain. The closed loop response of the loops 45a or 45 b introduces a cutoff frequency in the TIA transfer functionwhich equals the open loop GBW product (FC=GBW). The proposedarchitecture has several advantages over prior art topologies which are:

1) more convenient in realizing the low pass filter section 46 a and 46b (R_(C), C_(C)) than prior art, and

2) suitable for fully differential TIA topologies unlike prior art.

The actual realization of the R-C section of the low pass filters 46 aand 46 b is more convenient in the proposed architecture thanimplementing passive AC coupling circuitry at the TIA input as in priorart because of two reasons: 1) there is no upper limit on the maximumvalue of the resistor R_(C) as no DC current flows in it, and 2) C_(C)parasitic capacitance doesn't harm the TIA bandwidth as it is placedaway from the RF signal path between the GM cell input to the ground.

Furthermore, the proposed architecture is more suitable for fullydifferential front-end TIA architecture than the prior art. In fullydifferential TIAs 42, each output depends on both inputs due to the highcommon mode rejection of the fully differential TIA 42. Assuming anideal differential amplifier A_(O) employed in the TIA 42 (common modegain=0, differential mode gain=∞), TIA output voltages (V_(OUTP),V_(OUIN)) are expressed as,

$\begin{matrix}{{V_{OUTP} = {{- \frac{R_{F}}{2}}\left( {I_{P} - I_{N}} \right)}},} & {6a} \\{{V_{OUTN} = {\frac{R_{F}}{2}\left( {I_{P} - I_{N}} \right)}},} & {6b}\end{matrix}$

where I_(P) and I_(N) are the input positive and negative currents ofthe differential TIA 42, respectively. Equation (6) implies that bothV_(OUTP and V) _(OUTN) depend on I_(P) and I_(N) with the same weightand opposite effect. Thus, the prior art cannot be employed with a fullydifferential TIA because the two cancellation loops 45 a and 45 b willbe strongly coupled and affected by each other. However, in the pseudodifferential topology of the present invention, each output V_(OUTP) andV_(OUTN) (positive or negative) depends only on the corresponding inputcurrent I_(AC) which makes the two cancellation loops 45 a and 45 bdecoupled and the cancellation performed correctly. Accordingly, theproposed AC coupling scheme offers better isolation between the DCcancellation loops 45 a and 45 b, in particular with the fullydifferential TIA 42, because the sensing operation is performed at theinput of the TIA 42.

One way to implement the trans-conductance (G_(M)) cell with the looppass filter 45 a and 45 b R-C section for a single-ended TIA isillustrated in FIG. 5 for the proposed active AC coupling circuitry 40.The trans-conductance cell G_(M) draws a current (I_(OUT)) proportionalto the DC component of the differential input (V_(INP)−V_(INN)). The GMcell comprises a voltage amplifier 51 followed by a differential pair(T₁, T₂) of transistors. A current steering is performed linearly in thedifferential pair T1 and T2 using the differential input DC voltage(V_(INP)−V_(INN)) such that the output current equals I_(B)/2*N atVINP=VINN and its maximum value is I_(B)*N. The AC trans-conductance ofthe proposed GM cell is expressed as,

G_(N)=A×g_(m)×N₁

where A is the gain of the amplifier 51 and g_(m) is thetrans-conductance of the differential pair (T₁, T₂) each transistorhaving a resistor R connected thereto. N is a scaling factor for theoutput emitter-degenerated current mirror 52, where the output bipolartransistor 55 is made N longer and the degeneration resistor 56 is Ntimes smaller. N may or may not be an integer. The scaling factor Nenables the trans-conductance cell to operate with a smaller biascurrent from bias current source I_(B), and thereby to reduce powerdissipation. The bias current source I_(B) is connected to both of thetransistors T₁ and T₂ of the differential pair. The output I_(OUT) isconnected to the second transistor T₂ via an output transistor 55 andthe output resistor 56 R/N.

Accordingly, to determine the value for N, the first step is todetermine the value for I_(OUT) or V_(REF), based on systemrequirements, i.e. how much DC current I_(DC) needs to be cancelled. Forexample, if I_(OUT) is 4 mA, it is undesirable to burn another 4 mA inthe bias current I_(B). Accordingly, a scaling factor of, e.g. pickN=40, is selected, whereby IB=I_(OUT)/N=100 uA, which adds only a smallnumber to the overall power dissipation. In practice, N should not betoo large, but typically does not have to be, because at some point, itdoes not make sense to push N to much higher values, as the goal ofreducing power dissipation has typically already been achieved.Accordingly, N may be in the range from N=1 to N=1000, preferably N=4 toN=40, but it can be any another number, equal or greater than 1, N>1.Formally, there is no reason why N cannot be less than 1, but may bewasteful.

The capacitor C_(C) of the R_(C) loop pass filter section 46 a and 46 bmay be implemented using a miller capacitor between the input and outputof the voltage amplifier 51 to boost its value by the voltage amplifiergain, which helps in reducing the size of the implemented R-C section 46a and 46 b. The loop pass filter cutoff frequency is expressed as,

$f_{c} - {\frac{1}{A} \times {\frac{1}{2\pi \; R_{c}C_{c}}.}}$

There are several reasons why a fully differential TIA front end isbetter than the most commonly used single-ended. First, the fullydifferential TIA front end has an excellent CMRR (common mode rejectionratio). Coherent receivers often have challenging CMRR requirements, anda fully differential TIA easily meets and exceeds most CMRR specs.Second, fully differential TIA front ends have better power supply andground noise rejection. Third, the output of a fully differential TIA isfully compatible with the differential amplifier stages, so there is noneed to convert the signal from single-ended to differential.

However, the present invention may be used in non-coherent systems,since large DC currents can result from poor extinction ratio of theoptical signal or if APDs are used there could be a very significantdark current.

With reference to FIG. 6, an optical receiver front-end is composed of aphoto diode (PD) 61 followed by a trans-impedance amplifier (TIA) 62 andmain amplifiers (MAs) 63. The PD 61 receives a transmitted opticalsignal 64 and generates a current 66 proportional to the receivedoptical power of the received optical signal 64. The ratio between PDoutput current 6 to the input optical power of the optical signal 64 isthe photo diode responsivity (R). The TIA 62 converts the PD current 66to voltage, which is and then amplified by MAs 63 to the desired signallevel.

An active AC coupling circuitry is disposed between the photo diode (PD)61 and the TIA 62. An analog DC cancellation loop 67 is located prior toeach input of the TIA 62, and draws the photo diode DC current I_(DC),whereby only the AC signal I_(AC) is coupled to the TIA 62. Thecancellation loop 67 comprises a trans-conductance (GM) cell 68, and alow pass filter section 69, e.g. comprised of a cancellation resistorR_(C) and a cancellation capacitor C_(C). The input and output of thecancellation loop 67 is connected to the input of the TIA 62, i.e. boththe input and output of the loop 67 are between the PD 61 and the TIA62, as shown in FIG. 6. Photo diode DC current I_(DC) is drawn from theinput node of the TIA 62 in the GM cell 68, such that the analog loop 67maintains the DC voltage value of the TIA input node to be the same as areference voltage (V_(REF)). This reference voltage V_(REF) is an inputnode voltage of a replica TIA, which represent the actual TIA input nodevoltage for a zero DC current condition.

The foregoing description of one or more embodiments of the inventionhas been presented for the purposes of illustration and description. Itis not intended to be exhaustive or to limit the invention to theprecise form disclosed. Many modifications and variations are possiblein light of the above teaching. It is intended that the scope of theinvention be limited not by this detailed description, but rather by theclaims appended hereto.

1. An optical receiver comprising: a first photodetector (PD) forconverting a first input optical signal into a first PD currentcomprising a first AC component and a first DC component; atransimpendance amplifier (TIA) for converting the first AC componentinto a first voltage signal, the TIA comprising a first input; a signalpath electrically connecting the TIA to the first PD in the absence of acapacitor therebetween; and a first DC cancellation loop connected tothe signal path between the first PD and the TIA for cancelling thefirst DC component, the first DC cancellation loop comprising: a firstinput and a first output each connected to the first input of the TIA; afirst trans-conductance cell capable of drawing in the first DCcomponent, such that the first DC cancellation loop maintains a first DCvoltage value of the first input of the TIA substantially the same as afirst reference voltage V_(REF1), wherein V_(REF1) is equal to an actualTIA input voltage for a zero DC current condition; and a first low passfilter.
 2. The optical receiver according to claim 1, further comprisinga circuit for generating the first reference voltage.
 3. The opticalreceiver according to claim 1, further comprising a replica TIAconfigured to provide the first reference voltage V_(REF1).
 4. Theoptical receiver according to claim 1, wherein the first low pass filtercomprises a first resistor and a first capacitor; and wherein the firstcapacitor is placed away from the signal path between the TIA and thefirst PD.
 5. The optical receiver according to claim 4, wherein thefirst DC cancellation loop introduces a cutoff frequency in a transferfunction of the TIA, wherein the cut-off frequency equals an open loopgain bandwidth (GBW) product; wherein[[GBW=G_(M)×R_(F)/A_(o)×1/2πR_(C)C_(C)]]${{GBW} = {\frac{G_{M} \cdot R_{F}}{A_{O}} \cdot \frac{1}{2\pi \; R_{C}C_{C}}}};$wherein G_(M) is an AC trans-conductance of the first trans-conductancecell, R_(F) is resistance of a TIA feedback resistor, A_(O) is a TIAfeed-forward amplifier DC gain, R_(C) is a resistance of the firstresistor, and C_(C) is a capacitance of the first capacitor.
 6. Theoptical receiver according to claim 4, wherein the firsttrans-conductance cell comprises a voltage amplifier, a bias currentsource, a differential pair (T₁, T₂) of transistors, and a currentmirror; wherein the AC trans-conductance G_(M) of the firsttrans-conductance cell is defined by:[[G _(M) =A×g _(m) ×N ₁ ]]G _(M) =A·g _(m) ·N wherein A is the gain ofthe voltage amplifier, g_(m) is the trans-conductance of thedifferential pair (T₁, T₂), and N is a scaling factor of the currentmirror and wherein N≥1.
 7. The optical receiver according to claim 6,wherein the first capacitor comprises a miller capacitor disposedbetween the input and output of the voltage amplifier to boost the firstcapacitor's value by the gain A of the voltage amplifier.
 8. The opticalreceiver according to claim 6, wherein N is between 4 and 40 enablingthe trans-conductance cell to operate with a smaller bias current I_(B),to reduce power dissipation.
 9. The optical receiver according to claim1, wherein the first DC cancellation loop is configured to have an upperspeed limit, which is equal or smaller than a lowest frequency componentof the received optical signal, and a lower speed limit, which is equalor greater than a fastest variation of the DC component.
 10. The opticalreceiver according to claim 1, further comprising: a secondphotodetector (PD) for converting a second input optical signal into asecond PD current comprising a second AC component and a second DCcomponent; wherein the TIA comprises a differential TIA for furtherconverting the second AC component into a second voltage signal; and asecond DC cancellation loop between the second PD and the TIA forcancelling the second DC component, the second DC cancellation loopcomprising: a second input and a first second output, each DC connectedto an input of the TIA; a second trans-conductance cell (G_(M)), capableof drawing in the second DC component, such that the second DCcancellation loop maintains a second DC voltage value of the secondinput of the TIA the same as a second reference voltage V_(REF2)),wherein V_(REF1) is defined by an actual TIA input voltage for zero DCcurrent condition; and a second low pass filter.
 11. The opticalreceiver according to claim 10, further comprising: an input port forinputting a combined optical signal; a polarization beam splitter forsplitting combined optical signal into first and second polarizedcomponents; a local oscillator for generating first and secondoscillator components; a first hybrid mixer for generating first andsecond phase differentiate optical signals from the first polarizedcomponent and the first oscillator component; a second hybrid mixer forgenerating third and fourth phase differentiated optical signals fromthe second polarized component and the second oscillator component;wherein the first and second input optical signals comprise the firstand second phase differentiated optical signals.